In order to assist understanding of the invention as explained in the following text, the fundamental design of an electronic ballast, which is used to drive a florescent lamp, and its method of operation will first of all be explained with reference to FIGS. 1 and 2. A ballast such as this is described, by way of example, in EP 1 066 739 B1, U.S. Pat. No. 6,617,805 B2 or in the Data Sheet No. PD 601182-I for the IR2156 (S) integrated module produced by International Rectifier, California, USA.
An electronic ballast has a half-bridge with two semiconductor switching elements Q1, Q2, whose load paths are connected in series between supply terminals K1, K2, between which a DC voltage Vb is applied. These two semiconductor switching elements S1, S2 are driven via a drive circuit 20 which drives each of the two semiconductor switching elements S1, S2 in a clocked form. The two semiconductor switches Q1, Q2 are in this case driven alternately in order to ensure that the two semiconductor switches are never switched on at the same time. A voltage V2, which has an essential square waveform, is produced at an output K3 of the half-bridge, which is formed by a node that is common to the load paths of the semiconductor switching elements.
This voltage V2 feeds a resonant tuned circuit with a resonant inductance L1 and a resonant capacitor C1, with a florescent lamp being connected in parallel with the resonant capacitor C1 in the example. A further capacitor C2, which is connected in series with the resonant inductance L1 and upstream of the parallel circuit formed by the florescent lamp 10 and the resonant capacitor C1, is used as a blocking capacitor, and blocks direct-current components.
A snubber capacitor C3 is connected in parallel with the load path of the second semiconductor switching element Q2, with the object of reducing the switching losses during zero-voltage switching operation (ZVS) of the two semiconductor switching elements Q1, Q2.
The illustration does not show the normally provided measurement connections of the drive circuit 20, via which by way of example a voltage across the florescent lamp 10 or a current through the half-bridge Q1, Q2 is determined, and supply connections via which a voltage supply is provided for the drive circuit 20. The DC voltage Vb for the ballast is provided, for example, by a switched-mode converter with a power factor correction function (power factor controller, PFC). In this context, reference is made, for example, to EP 1 066 739 B1, U.S. Pat. Nos. 6,617,805 B2 or 6,400,095 B1, as cited above.
FIG. 2 shows the waveform of the output voltage V2, which is produced between the output terminal K3 of the half-bridge Q1, Q2 and the reference ground potential GND, of the half-bridge circuit Q1, Q2, of the current Iq2 through the second semiconductor switching element Q2, the current I1 into the load that is connected to the half-bridge circuit Q1, Q2, and the drive signals S1, S2 for the semiconductor switching elements S1, S2 for a disturbance-free operating state after starting of the florescent lamp.
The semiconductor switching elements Q1, Q2 are switched on by the drive circuit 20 via the drive signals S1, S2, with a respective phase shift, for switched-on durations Ton1, Ton2, with the drive periods Tp for the two semiconductor switches S1, S2 each being the same. The drive is provided, for example, in such a way that there is a minimum switched-off time toff between one of the two semiconductor switching elements being switched off and the other being switched on. The switched-on durations Ton1, Ton2 are normally each of equal length, the duty cycle, that is to say the ratio of the switched-on duration to the period duration is, for example, about 45%.
When the first semiconductor switch S1 is switched on and the second semiconductor switch S2 is switched off, the output voltage V2 from the half-bridge circuit Q1, Q2 corresponds approximately to the DC voltage Vb between the terminals K1, K2, ignoring the switched-on resistance of the first semiconductor switching element Q1. This voltage results in a lamp current I1, which flows in the opposite direction to that shown in FIG. 1 and whose magnitude increases as the time for which the first semiconductor switching element S1 is switched on increases. Once the first semiconductor switching element S1 has been switched off, this current is first of all still maintained by virtue of the inductance L1 of the series tuned circuit L1, C1 and thus discharges the snubber capacitor C3, which is connected in parallel with the second semiconductor switching element Q2, as a result of which the voltage across the load path of this second semiconductor switching element Q2 tends to zero. Once this capacitor C3 has been discharged, the body diode of the second semiconductor switching element Q2, which is in the form of an n-channel MOSFET, carries the lamp current I1, in this case acting as a freewheeling diode. This lamp current I1 changes its polarity in the time period after the second semiconductor switching element S2 has been switched on, and flows in the direction shown in FIG. 1 before the second semiconductor switching element S2 is switched off. Once the second semiconductor switching element Q2 has been switched off, the snubber capacitor C3 is charged via the current flowing through the inductance L1 to the value of the DC voltage Vb, with any further voltage rise being limited by an integrated body diode in the first semiconductor switching element, which is formed by an n-channel MOSFET. In this case, the first semiconductor switching element Q1 is not switched on until the voltage at the output K3 has risen to the value of the DC voltage Vb, and the voltage across the load path of the first semiconductor switching element, Q1 is thus zero.
The snubber capacitor C3 assists zero-voltage switching of the first and second semiconductor switching elements Q1, Q2, that is to say switching of these semiconductor switching elements Q1, Q2 when the voltage across their load path is equal to zero. The switches Q1, Q2 can admittedly also be switched on at zero voltage without the snubber capacitor C3. The only precondition for this is that the current through the load path continues to flow with the same polarity until the corresponding switch Q1, Q2 is switched on. Without any snubber capacitor C3, the voltage would, however, rise very quickly after switching off a switch Q1, Q2, leading to corresponding switching-off losses. The snubber capacitor C3 limits this rate of voltage rise, and thus reduces the switching losses.
However, situations in which such zero-voltage switching operation cannot be achieved may occur during operation of a florescent lamp. In this case, the snubber capacitor C3 charge is not changed by means of the current that is induced in the resonant inductance L1 but by means of the currents flowing through the semiconductor switching elements on switching on, and this is associated with considerable losses. Operating states such as these may occur, for example, when the lamp has been removed from the socket or is damaged, or when the DC voltage Vb falls for a lengthy time period during normal operation.
In order to avoid overloading of the semiconductor switching elements which are designed to be continuously loaded only for zero-voltage switching operation during non-zero-voltage switching operation, it is necessary to identify an operating state such as this and, if necessary, to switch off the florescent lamp by interrupting the drive to the half-bridge if this operating state lasts for longer than a predetermined time period.
In order to detect such non-zero-voltage switching operation, it is known from U.S. Pat. Nos. 6,331,755 B1 and 5,973,943 for a current to be detected by the low-side switch in the half-bridge and to be assessed against a reference value at the time at which the switch is switched on and off. U.S. Pat. No. 6,400,095 B1 and EP 1 066 739 B1 propose that the current through the lamp be detected by means of a shunt resistance, and be assessed against a reference value.